HIP6018B 데이터 시트보기 (PDF) - Intersil

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HIP6018B

Advanced PWM and Dual Linear Power Control

Intersil 

HIP6018B

LUV

OC1

0.15V +

-

SS

+

4V -

OV

OVER

CURRENT

LATCH

SQ

INHIBIT

R

S

COUNTER

R

FAULT VCC

LATCH

UP

SQ

POR

R

FAULT

FIGURE 7. FAULT LOGIC - SIMPLIFIED SCHEMATIC

and VOUT3 to slew up over increased time intervals, without

generating a fault. Cycling the bias input voltage (+12VIN on

the VCC pin) off then on resets the counter and the fault latch.

Over-Voltage Protection

During operation, a short on the upper PWM MOSFET (Q1)

causes VOUT1 to increase. When the output exceeds the

over-voltage threshold of 115% (typical) of DACOUT, the

over-voltage comparator trips to set the fault latch and turns

Q2 on as required in order to regulate VOUT1 to 1.15 x

DACOUT. This blows the input fuse and reduces VOUT1.

The fault latch raises the FAULT pin close to VCC potential.

A separate over-voltage circuit provides protection during

the initial application of power. For voltages on the VCC pin

below the power-on reset (and above ~4V), VOUT1 is

monitored for voltages exceeding 1.26V. Should VSEN1

exceed this level, the lower MOSFET (Q2) is driven on as

needed to regulate VOUT1 to 1.26V.

Over-Current Protection

All outputs are protected against excessive over-currents.

The PWM controller uses the upper MOSFET’s on-

resistance, rDS(ON) to monitor the current for protection

against shorted outputs. The linear regulator monitors the

current of the integrated power device and signals an over-

current condition for currents in excess of 230mA.

Additionally, both the linear regulator and the linear

controller monitor FB2 and FB3 for under-voltage to protect

against excessive currents.

Figures 8 and 9 illustrate the over-current protection with an

overload on OUT1. The overload is applied at T0 and the

current increases through the output inductor (LOUT1). At time

T1, the OVER-CURRENT1 comparator trips when the voltage

across Q1 (ID • rDS(ON)) exceeds the level programmed by

ROCSET. This inhibits all outputs, discharges the soft-start

capacitor (CSS) with a 11μA current sink, and increments the

counter.CSS recharges at T2 and initiates a soft-start cycle

with the error amplifiers clamped by soft-start. With OUT1 still

overloaded, the inductor current increases to trip the over-

current comparator. Again, this inhibits all outputs, but the

soft-start voltage continues increasing to 4V before

discharging. The counter increments to 2. The soft-start cycle

repeats at T3 and trips the over-current comparator. The SS

pin voltage increases to 4V at T4 and the counter increments to

3. This sets the fault latch to disable the converter. The fault is

reported on the FAULT pin.

FAULT

10V

REPORTED

0V

COUNT

=1

4V

COUNT

=2

COUNT

=3

2V

0V

OVERLOAD

APPLIED

0A

T0 T1

T2

TIME

T3

T4

FIGURE 8. OVER-CURRENT OPERATION

The linear regulator operates in the same way as PWM1 to

over-current faults. Additionally, the linear regulator and

linear controller monitor the feedback pins for an under-

voltage. Should excessive currents cause FB2 or FB3 to fall

below the linear under-voltage threshold, the LUV signal

sets the over-current latch if CSS is fully charged. Blanking the

LUV signal during the CSS charge interval allows the linear

outputs to build above the under-voltage threshold during

normal start-up. Cycling the bias input power off then on

resets the counter and the fault latch.

Resistor ROCSET1 programs the over-current trip level for the

PWM converter. As shown in Figure 9, the internal 200μA

current sink develops a voltage across ROCSET (VSET) that is

referenced to VIN. The DRIVE signal enables the over-current

comparator (OVER-CURRENT1). When the voltage across the

upper MOSFET (VDS(ON)) exceeds VSET, the over-current

comparator trips to set the over-current latch. Both VSET and

VDS are referenced to VIN and a small capacitor across

ROCSET helps VOCSET track the variations of VIN due to

MOSFET switching. The over-current function will trip at a peak

inductor current (IPEAK) determined by:

IPEAK

=

I--O-----C-----S-----E----T----x----R----O------C----S-----E----T--

rDS(ON)

8

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