TSM108 데이터 시트보기 (PDF) - STMicroelectronics

부품명

상세내역

제조사

TSM108

VOLTAGE AND CURRENT STEPDOWN PWM CONTROLLER

STMicroelectronics 

TSM108

9.1. Inductor

The minimum inductor value to choose should ap-

ply to

Lmin = (1 - D) R / 2F

where R = Vout / Iout = 9.6W

and where D = Vout / Vin = 0.5

Therefore, Lmin = 24µH.

The frequency may vary depending on the tem-

perature, due to the fact that the frequency is fixed

by an external capacitor. Therefore, we must cal-

culate the inductor value considering the worst

case condition in order to avoid the saturation of

the inductor, which is when the battery voltage is

at it's highest, and the switching frequency at it's

lowest. Thanks to the OVLO onboard function, the

operation of the DC/DC converter will be stopped

as soon as the voltage exceeds the OVLO level.

Let's suppose the OVLO pin has been left open,

therefore, the maximum input voltage of the DC/

DC converter will be Vin max. = 32V. Frequency

min stands in the range of 75kHz

In this case, D = 6 / 32 = 0.1875, therefore Lmin =

52µH.

If we allow a 25% security margin

Lmin = 68µH

9.2. Capacitor

The capacitor choice will depend mainly on the ac-

cepted voltage ripple on the output

Ripple = DVout / Vout = (1-D) / 8LCF²

Therefore, C = (1-D) / 8LRippleF². If C = 22µF,

then Ripple = 0.4% which should be far accept-

able.

Here again, the worst conditions for the ripple are

set when the input voltage is at the highest (32V)

and the frequency at it's lowest (75kHz).

with C = 22µF, Ripple = 1.2%

9.3. Ratings for the Inductor, Capacitor,

Transistor and Diode

The inductor wire must be rated at the rms current,

and the core should not saturate for peak inductor

current. The capacitor must be selected to limit the

output ripple to the design specifications, to with-

stand peak output voltage, and to carry the re-

quired rms current.

The transistor and the diode should be rated for

the maximum input voltage (up to 60V). The recir-

culation diode has to be a Schottky type for effi-

ciency maximization or ultrafast recovery.

A compromise between the switching and conduc-

tion losses of the external power element has to

be found.

Losses in the switch are:

Pswitch = Prise + Pfall + Pon

where Prise + Pfall represent the switching losses

and where Pon represents the conduction losses.

Prise + Pfall = Iout x Vin x (Trise + Tfall) x F / 2

Pon = Ron x Iout² x d

where Trise is the switching on time, and Tfall is

the switching off time, and where d is the duty cy-

cle of the switching profile, which can be approxi-

mated to 1 under full load conditions.

With the two last equations, we can see easily that

what we may gain by choosing a performing low

Rdson Pchannel MOSFET (for example) may be

jeopardised by the long on and off switching times

required when using a large input gate capaci-

tance.

10. Electromagnetic Compatibility

The small schematic hereafter shows how to re-

duce the EMC noise when used in an EMC sensi-

tive environment:

EMC Improvement

MOSFET P

Q1

D1 L1

GD

TSM108

The RC components should realise a time con-

stant corresponding to one tenth of the switching

time constant of the TSM108 (i.e. in our example,

the oscillator frequency is set to 10µs correspond-

ing to 100kHz, therefore, the RC couple should re-

alise a time constant close to 1µs).

Choosing the components must privilege a rather

small resistivity (between 10 to 100W). A guess

couple of values for RC in our example would be:

R= 22W, C= 47nF

11. Efficiency Calculations (rough estimation)

The following gives a rough estimation of the effi-

ciency of a car phone charger, knowing that the

exact calculations depend on a lot of parameters,

as well as on a wide choice of external compo-

nents.

Let’s consider the following characteristics of a

classical car phone charger application:

❑ Vin = Vcc = 12V, Iout = 625mA, Vout = 6V

9/13

Share Link: